Multipath discriminator module for a navigation system

ABSTRACT

A multipath discriminator module and related methods are provided for communications and/or navigation systems that implement spread spectrum modulation. In one example embodiment, the module includes an input suitable for receiving navigation signals, a sampler for supplying sampled signals at a frequency twice the apparent frequency of the code of the signals, and a submodule for calculating an error signal from the sampled signals.

The invention relates to a multipath discriminator module for anavigation system, and also to a navigation system including such amodule.

BACKGROUND OF THE INVENTION

Over the last ten years, direct sequence and spread spectrum (DS-SS)modulation systems have been increasing in importance.

At present, this technique is implemented not only in satellitenavigation systems such as GPS and GLONASS, but it has also beenintroduced into terrestrial and satellite communications systems, e.g.US standard IS-95, GLOBALSTAR, and more recently in the third generationof mobile telephones using the UMTS standard, and also in the Europeansatellite navigation system GALILEO.

The concept of DS-SS modulation, e.g. bi-binary phase shift keying(Bi-BPSK) introduces a pseudo-random noise (PRN) code which has theconsequence of the resulting modulated signal presenting a passband thatis wider than a signal that is transmitting only the data signals. It isin this sense that the spectrum density of the signal is said to be“spread”.

In the receiver, a locally-generated replica of the transmitted PRN codeis aligned with the phase of the code of the received signal. Inparticular, in navigation receivers, code phase alignment is essentialfor determining accurately the time of arrival (TOA), which is used fordetermining the geometrical distance between the transmitter and thereceiver. Once alignment has been achieved, it is possible to estimatecarrier phase and to determine the symbols of the transmitted data.

This alignment is conventionally achieved in the receiver by means of adelay-locked loop (DLL), an example of which is described in the articleby M. Simon et al. published in the work “Spread spectrum communicationshandbook” published by McGrawHill, Inc., 2nd edition, 1994.

Such alignment uses the result of correlation between the receivedsignal and early and late versions (E and L) of a locally-generatedreference code signal in order to calculate an error signal that isproportional to code phase error (the difference between the estimatedcode phase and the received phase).

This error signal must indicate the direction in which the phase of thereference signal needs to be offset (advanced or retarded) in order tobe brought into synchronization with the received signal. The spacingbetween the early and late codes (E and L) is generally one bit of apseudo-noise sequence known as a “chip”.

Signals of the square-root raised cosine (SRC) type (which have a raisedcosine spectrum) are defined in the UMTS standard. The same type of SRCsignal is likely also to be adopted in the above-mentioned GALILEOsystem. A digital receiver implementing such signals is described in thearticle by R. de Gaudezni et al. entitled “A digital chip timingrecovery loop for band-limited direct-sequence spread-spectrum signals”published in IEEE Trans. on Comm., Vol. 41, No. 11, pp. 1760-1769,November 1993.

The accuracy with which time of arrival is measured is negativelydisturbed by the presence of distortion due to multiple paths, and as aresult, when performing telemetry, the precision with which position isdetermined is decreased, and when transmitting data, there is anincrease in bit or frame error rate. This is particularly true when themultipath distortion is represented essentially by a single reflectioncoming from a point which is situated in the immediate environment ofthe receiver with a small dynamic range.

The superposition of the direct and reflected signals is thus liable togive rise to jitter which affects the TOA measurements performed by theDLL.

As a result, techniques that make it possible to reduce the impact ofmultiple paths on determining code phase are of very great interest,particularly in the field of navigation.

Until now, methods for compensating multiple paths for use in telemetryhave been developed essentially in the context of GPS receivers.

As a result, a large number of those algorithms make use of the factthat the apparent chip rate of the publically available C/A code is muchlower than the passband of the transmission. It is then advantageous toreduce the time differences between the early and late reference codesignals E and L until they have a value that is less than one bit of apseudo-random noise sequence (“one chip duration”) in order to reducethe error that is induced by the multipath beams.

In the context of SRC type systems, given that the frequency spectrum isstrictly limited to (1+β) times the apparent code rate (where βdesignates the attenuation factor of SRC pulses), the above-mentionedmethods are not effective in compensating multipath beams.

In the article by Philip G. Mattos entitled “Multipath elimination forthe low-cost consumer GPS” published in the Proceedings of the ION GPS1996 Conference in Kansas City, pp. 665-671, it has also been suggestedto replace the early and late correlation points E and L by two earlycorrelation points. However, that article does not give any means forimplementing that technique.

OBJECTS AND SUMMARY OF THE INVENTION

An object of the present invention is to provide a discriminator modulewhich is suitable for use in a spread spectrum communications ornavigation system, and more particularly one using SRC type modulation.

In a first variant, the invention provides a multipath discriminatormodule for a communications and/or navigation system that implementsspread spectrum modulation, which module has an input suitable forreceiving navigation signals, a sampler for supplying sampled signals ata frequency twice the apparent frequency fc of the code of said signals,and a submodule for calculating an error signal e_(k) from said sampledsignals, and a locally generated spreading code C|K|L wherein:

$e_{k} = {K_{\beta}\mspace{14mu}{\Re( {\frac{Z_{K}\text{--}}{Z_{K} -} + {S\;\beta}} )}}$${with}\mspace{14mu}\{ {\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k - {t\; 2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k - {t\; 1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}{or}\mspace{14mu}\{ {{\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}K_{\beta}} = {{{constant}\mspace{34mu}{and}S_{\beta}} = {{{- \frac{g( {{- t}\; 1{Tc}} )}{g( {{- t}\; 2{Tc}} )}}\mspace{40mu}{with}{g({aTc})}} = \frac{{Sin}\;\Pi\;\alpha\;{Cos}\;\Pi\;\beta\;\alpha}{\pi\;{\alpha\lbrack {1 - ( {2\beta\; a} )} \rbrack}}}}} } $β designating the attenuation factor of the SRC signal.

In a second variant, the invention provides a multipath discriminatormodule for a communications and/or navigation system that implementsspread spectrum modulation, which module has an input suitable forreceiving navigation signals, a sampler for supplying sampled signals ata frequency twice the apparent frequency fc of the code of said signals,and a submodule for calculating an error signal e_(k) from said sampledsignals, and a locally generated spreading code C|K|L wherein:

$e_{k} = {K_{\beta}\;( {{- \frac{Z_{K}\text{--}}{Z_{K} -}} + {S\;\beta}} )}$${with}\mspace{14mu}\{ {\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k - {t\; 2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k - {t\; 1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}{or}\mspace{14mu}\{ {{\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}K_{\beta}} = {{{constant}\mspace{45mu}{and}\mspace{14mu} S_{\beta}} = {{\frac{g( {{- t}\; 1{Tc}} )}{g( {{- t}\; 2{Tc}} )}\mspace{45mu}{with}{g({aTc})}} = \frac{{Sin}\;\Pi\;\alpha\;{Cos}\;\Pi\;\beta\;\alpha}{\pi\;{\alpha\lbrack {1 - ( {2\beta\; a} )} \rbrack}}}}} } $β designating the attenuation factor of the SRC signal.

In a third variant, the invention provides a multipath discriminatormodule for a communications and/or navigation system that implementsspread spectrum modulation, which module has an input suitable forreceiving navigation signals, a sampler for supplying sampled signals ata frequency twice the apparent frequency fc of the code of said signals,and a submodule for calculating an error signal e_(k) from said sampledsignals, and a locally generated spreading code C|K|L wherein:

$e_{k} = {K_{\beta}\mspace{20mu}{\mathcal{R}( {{2\frac{Z_{K}\text{--}}{Z_{K}^{-} + Z_{K}^{+}}} + S_{\beta}} )}}$${with}\mspace{14mu}\{ {\begin{matrix}{Z_{K}^{+} = {\lfloor {r_{k + 0.5} \cdot C_{{K}L}} \rfloor \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k - {t\; 2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{{kt} - {t\; 1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}{or}\mspace{14mu}\{ {{\begin{matrix}{Z_{K}^{+} = {\lbrack {r_{k} \cdot C_{{{k + {1/2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{({k - {t\; 2}})}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k} \cdot C_{{({k - {t\; 1}})}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}\beta} = {{{constant}\mspace{34mu}{and}\mspace{14mu} S_{\beta}} = {{{- \frac{g( {{- t}\; 1{Tc}} )}{g( {{- t}\; 2{Tc}} )}}\mspace{40mu}{with}\mspace{14mu}{g({aTc})}} = \frac{{Sin}\;\Pi\;\alpha\;{Cos}\;\Pi\;\beta\;\alpha}{\pi\;{\alpha\lbrack {1 - ( {2\beta\; a} )} \rbrack}}}}} } $β designating the attenuation factor of the SRC signal.

In each of the above cases, h_(k) ^(b) designates the impulse responseof a lowpass filter.

The discriminator module may be such that:

${K\;\beta} = \frac{1}{\frac{\mathbb{d}}{\mathbb{d}ɛ}( \frac{g( {( {ɛ - {t\; 1}} ){Tc}} )}{g( {( {ɛ - {t\; 2}} ){Tc}} )} )_{ɛ = 0}}$

In each of the three above variants, it is possible to have t1=1.5 andt2=0.5.

The invention also provides a multipath discriminator module for acommunications and/or navigation system that implements spread spectrummodulation, which module has an input suitable for receiving navigationsignals, a sampler for supplying sampled signals at a frequency twicethe apparent frequency fc of the code of said signals, and a submodulefor calculating an error signal e_(k) from said sampled signals, and alocally generated spreading code C|K|L wherein the submodule calculatesthe real portion of a ratio of two advance correlation values relativeto the real phase value, these values coming from correlation betweenthe received signal and the locally generated reference signal.

The invention also provides a navigation system, presenting adiscriminator module as defined above.

Finally, the invention provides a navigation system, presenting adiscriminator module generating an error signal e′_(k) serving inconventional manner to correct a closed loop on the basis of sampledsignal Z+_(K) and Z−_(K) which is associated with a discriminator moduleas defined above in order to operate in an open loop on a said errorsignal e_(k) to generate a correction signal for the code phase output.

BRIEF DESCRIPTION OF THE DRAWINGS

Other characteristics and advantages of the invention appear better onreading the following description given by way of non-limiting exampleand with reference to the accompanying drawings, in which:

FIG. 1 shows the influence of a multipath signal comprising a directpath and at least one reflected path;

FIG. 2 is a block diagram of a DLL as described in the above-mentionedarticle by R. de Gaudenzi et al.;

FIGS. 3 a and 3 b are graphs plotting the autocorrelation function g(t)respectively for a module in accordance with FIG. 2 and in accordancewith the invention;

FIG. 4 is a block diagram of a DLL incorporating a discriminator of theinvention; and

FIG. 5 is a block diagram of a DLL incorporating a conventionaldiscriminator and a discriminator module of the present invention.

MORE DETAILED DESCRIPTION

Formula 3-1 represents a direct sequence spread spectrum (DS-SS) signals_(T)(t) in baseband, with the spreading code word having a length of 2pseudo-noise sequence bits (or “chips”), and each data symbol dp,q,ipresenting M/L distributed code words:

$\begin{matrix}\begin{matrix}{{S_{T}(t)} = {\sqrt{\alpha \cdot P_{s}} \cdot {\sum\limits_{i = {- \infty}}^{\infty}\;{( {{{{d_{P}}_{\cdot}}_{{\lfloor i\rfloor}_{M}} \cdot c_{P \cdot {i}_{L}}} + {j \cdot b \cdot d_{Q \cdot {i}_{M}} \cdot c_{Q \cdot {i}_{L}}}} ) \cdot}}}} \\{g_{T}( {t - {iT}_{c}} )}\end{matrix} & ( {3\text{-}1} )\end{matrix}$

The factors a and b are as follows in the following circumstances:

-   a=1.0 and b=0 BPSK DS-SS-   a=0.5 and b=1, with d_(P.|i|) _(M) =d_(Q.|i|) _(M) QPN DS-SS-   a=0.5 and b=1, with d_(P.|i|) _(M) =d_(Q.|i|) _(M) Bi-BPSK DS-SS    with:-   Ps: transmitted power-   d_(P/Q,i) data symbols (d_(P/Q,iε[−1.1]))-   C_(P/Q,i) bit (“chip”) of a spreading code word of length L    (C_(P/Q,iε[−1.1]))-   T_(c)=1/f_(c) duration of one bit of a pseudo-noise sequence (or    duration of one “chip”)-   g_(T)(t) pulse shape of a bit or “chip” e.g. SRC-   M data symbol length in the duration of one bit or “chip”-   |i|_(M) int(i/M)-   |i|_(M) imodM

After transmission by a channel with added white Gaussian noise (AWGN)presenting symmetrical spectrum density N_(o/2), the filtered receivedsignal r(t) is given by formula (3-2).

$\begin{matrix}\begin{matrix}{{r(t)} = {{S_{T}( {t - \tau} )} \otimes {g_{r}(t)}}} \\{= {\sqrt{a \cdot P_{s}} \cdot {\sum\limits_{i = {- \infty}}^{\infty}{( {{d_{P \cdot {\lfloor i\rfloor}_{M}} \cdot c_{P \cdot {i}_{L}}} + {j \cdot b \cdot d_{Q \cdot {\lfloor i\rfloor}_{M}} \cdot C_{Q \cdot {i}_{L}}}} ) \cdot}}}} \\{{{g( {t - \tau - {iT}_{c}} )} \cdot {\exp( {j( {{\Delta\;\omega\;(t)} + {\phi\;(t)}} )} )}} + {n(t)}}\end{matrix} & ( {3\text{-}2} )\end{matrix}$

g(t)=g_(T)(t){circle around (x)} g_(T)(t) designating the pulse shape ofone bit or “chip” after filtering and {circle around (x)} designatingconvolution.

g(t) thus constitutes the autocorrelation function of g_(T)(t).

Δω(t) designates the residual difference of the carrier frequency afterthe signal has been mixed into baseband.

φ(t) designates the phase of the carrier phase of the received signal.

Because of the similarity between the in-phase and quadrature componentsof the signal given by formula 3-2, it is appropriate to conserve onlythe in-phase component below. As a result, the symbols p,q designatingthese two situations are not conserved, in order to make the notationmore readable.

Formula 3-2 thus becomes:

$\begin{matrix}\begin{matrix}{{r(t)} = {\sqrt{P_{s}} \cdot {\sum\limits_{i = {- \infty}}^{\infty}{d_{{\lfloor i\rfloor}_{M}} \cdot c_{{i}_{L}} \cdot {g( {t - \tau - {iT}_{c}} )} \cdot}}}} \\{{\exp( {j( {{\Delta\;\omega\;(t)} + {\phi(t)}} )} )} + {n(t)}}\end{matrix} & ( {3\text{-}3} )\end{matrix}$for a system of the BPSK DS-SS type, for example.

FIG. 1 is a model of multiple paths that is suitable for use in showingthe effect of multiple paths on the DLL. In addition to the directsignal coming from the satellite, the antenna also receives a delayedsecond version of the same signal, referred to as the multipathcomponent and due to reflection, with the delay in this second signalcoming from the fact that it has had to follow a path that is longer.

The sum of these two signals as received by the antenna can be expressedusing the following formula:S′ _(T)(t)=S _(T)(t−τ)+α·S _(T)(t−τ−Δτ)·exp(jφ)  (3-4)with

-   -   α: attenuation of the reflected signal relative to the direct        signal    -   Δτ: delay of the reflected signal relative to the direct signal    -   φ=2π·Δτ·fc/c: the phase offset of the carrier of the reflected        signal relative to the direct signal (c=speed of light,        fc=carrier frequency).

After filtering (see formulae 3-2 and 3-3), the following is obtained:r(t)=r(t)+α·r(t−Δτ)·exp(jφ)  (3-5)

The explanation is given taking account of one reflection only. Inpractice, there are several components corresponding to multiplereflection paths which are all superposed on the direct signal.

The above-mentioned document by R. de Gaudenzi et al. uses a DLL ofarchitecture outlined below, when describing FIG. 2.

It should be observed:

-   -   that the analog-to-digital converter could be located at a        different location; and    -   that the time offset for aligning the local reference signal        with the received signal may be applied either to the local        reference signal or to the received signal.

The baseband signal r(t) obtained by filtering using a filter ofcharacteristic G_(T)(f) is sampled at twice the bit or “chip” frequency,i.e. at 2 fc. The samples that correspond to half-integer instants(k+0.5) Tc+τ are directed to the DLL, whereas the other samplescorresponding to “integer” instants kTc+τ are directed to instantaneouscorrelation and tracking of carrier phase and data demodulation (circuitNCO).

The samples corresponding to half-integer instants are given by:

$\begin{matrix}\begin{matrix}{{r_{k} + {1\text{/}2}} = {{\sqrt{P_{S}} \cdot \exp}\;{( {j\;\phi} ) \cdot {\sum\limits_{i = {- \infty}}^{\infty}{d_{{\lfloor i\rfloor}_{M}} \cdot c_{{i}_{L}} \cdot}}}}} \\{g{( {( {ɛ_{k} + k + {1\text{/}2} - i} )T_{c}} ) \cdot {+ n_{k + {1/2}}}}}\end{matrix} & ( {3\text{-}6} )\end{matrix}$

These samples r_(k+1/2) are directed along two branches. In the upperbranch (FIG. 2), the samples are delayed by one bit or “chip” Tc priorto being multiplied by the k^(th) value of the spreading code C_(|k|)_(L) as locally generated by the spreading code generator SCGEN. Thismultiplication is followed in each of the branches by a lowpass filterH^(b)(z).

This produces the following samples:

$\begin{matrix}{{Z_{k}^{+} = {\lbrack {r_{k} + {{{1/2} \cdot c}{k}_{L}}} \rbrack \otimes h_{k}^{b}}}{Z_{k}^{-} = {\lbrack {r_{k} - {{{1/2} \cdot c}{k}_{L}}} \rbrack \otimes h_{k}^{b}}}} & \text{(3-7)}\end{matrix}$where:

-   -   h_(k) ^(b) designates the impulse response of the lowpass        filter. The passband of this filter is limited in practice on        the low side by:        -   the rate f_(s) (f_(s)=f_(c)/M) of data symbols, otherwise            useful energy is lost; or        -   the dynamic range between the transmitter and the receiver,            the distance between the transmitter and the receiver is            assumed to be constant relative to the passband of H^(b)(z).

The error signal e_(k) is generated as follows:e _(k)=|_(Zk) ⁻|²−|_(Zk) ⁺|²  (3-8)

In order to obtain an error signal that can be used directly by theoperational circuit NCO, the signal e_(k) is generally filtered byanother digital filter, the loop filter, whose transfer function isH^(d)(z). Given that e_(k) is independent of the data symbols, thecharacteristics of H^(d)(z) are determined mainly by the responsedesired of the DLL to the dynamic range between the transmitter and thereceiver, the phase estimator loop needing to be capable of tracking alinear distance between the transmitter and the receiver that isincreasing or decreasing, without any residual tracking error.

The characteristic n(ε) indicates how the error signal e_(k) depends onthe code phase error ε.n(ε)=E[e _(k)|ε_(k) =ε∀k]  (3.9)

E [•] designating probability.

This gives:n(ε)=g ²[(ε−0.5)T _(c) ]−g ²[(ε+0.5)T _(c)]  (3.10)with SRC coding having an attenuation factor β, g(εT_(c)) becomes:

${g( {ɛ\; T_{c}} )} = {\frac{\sin( {\Pi\; ɛ} )}{\pi\; ɛ} \cdot \frac{\cos( {\pi\;\beta\; ɛ} )}{1 - ( {2\;\beta\; ɛ} )^{2}}}$The autocorrelation function g(εT_(c)) with β=0.35 is shown in FIG. 3 a.The early and late samples E and L of formula (3-8) are given for ε=0(no code phase error).

According to formula (3-10), the resulting S-shaped curve is shown inFIG. 3 a. Using this curve, the DLL controls the interpolator so thatthe received signal is in alignment with the locally-generated signal(i.e. ε=0).

The discriminator of the invention (FIG. 4) implements two early samplesE1(Z_(k) ⁻) and E2(Z_(k) ⁻⁻) at instants (e−t₁)T_(c) and (ε−t₂)T_(c),generated using the formula:

$\begin{matrix}{e_{k} = {{\kappa_{\beta} \cdot {\mathcal{R}( {\frac{{Zk}^{--}}{{Zk} -} + S_{\beta}} )}\mspace{79mu} \cdot {\mathcal{R}(\bullet)}}\mspace{14mu}{designating}\mspace{14mu}{the}\mspace{14mu}{real}\mspace{14mu}{{portion}\mspace{79mu} \cdot K_{\beta}}\mspace{14mu}{is}\mspace{11mu} a\mspace{14mu}{{constant}\mspace{79mu} \cdot S_{\beta}}\mspace{14mu}{is}\mspace{14mu}{an}\mspace{14mu}{{offset}\mspace{79mu} \cdot {the}}\mspace{14mu}{two}\mspace{11mu}{samples}\mspace{14mu} E\; 2\mspace{11mu}{and}\mspace{14mu} E\; 1\mspace{14mu}{are}\mspace{14mu}{defined}}} & ( {{see}\;( {3\text{-}7} )} )\end{matrix}$by:Z _(k) ⁻ =└r _(k−t2) ·C _(P,|K|L) ┘{circle around (x)}h _(k) ^(b) etZ_(k) ⁻⁻ =└r _(k−t1) ·C _(P,|K|L) ┘{circle around (x)}h _(k) ^(b)with, for example, t₁=1.5 and t₂=0.5.

It is also possible to generate these two samples by declaring the localreplica of the code using the formula:Z _(K) ⁻ =└r _(k) ·C _(P,|(k−t) ₂ _()|) L┘{circle around (x)}h _(k) ^(b)andZ _(K) ⁻⁻ =└r _(k) ·C _(P,|(k−t) ₁ _()|L) ┘{circle around (x)}h _(k)^(b)

The offset should be selected so that the expected value of the errorsignal E[e_(k)] is zero when the code phase error ε_(k) is zero.

Applying formula (3-12), this gives:

$\begin{matrix}{{E\lbrack { {{{\mathcal{R}( {\frac{Z_{\kappa}^{--}}{Z_{\kappa}^{-}} + S_{\beta}} )} {ɛ_{k} = 0} \rbrack} \equiv 0}\Rightarrow{S\;\beta}  = {{{E\lbrack {- {\mathcal{R}( \frac{Z_{\kappa}^{--}}{Z_{\kappa}^{-}} )}} }ɛ_{k}} = 0}} \rbrack} = {- \frac{g( {{- t}\; 1{Tc}} )}{g( {{- t}\; 2{Tc}} )}}} & ( {3\text{-}13} )\end{matrix}$

The slope factor K_(β) is preferably selected so that the value of theslope:

$E\lbrack {{\frac{\mathbb{d}\;}{\mathbb{d}ɛ}e_{k}{ɛ_{k}}} = 0} \rbrack$is equal to 1 when ε_(K)=0This gives:

$\begin{matrix}{K_{\beta} = {\frac{1}{E\lbrack {\frac{\mathbb{d}\;}{\mathbb{d}ɛ}{\mathcal{R}( {\frac{Z_{\kappa}^{--}}{Z_{\kappa}^{-}} + S_{\beta}} )} {ɛ_{k} = 0} \rbrack} } = \frac{1}{{\frac{\mathbb{d}\;}{\mathbb{d}ɛ}( \frac{g( {( {ɛ - t_{1}} ){Tc}} )}{g( {( {ɛ - {t\; 2}} ){Tc}} )} )ɛ} = 0}}} & (3.14)\end{matrix}$for example with t₁=1.5 and t₂=0.5.

As shown by formula (3-12), the discriminator e_(K) of the invention isindependent of the phase Φ of the carrier. This dependence is eliminatedby generating the ratio Z_(K) ⁻⁻/Z_(K) ⁻.

The autocorrelation function g(t) for β=0.35 is given in FIG. 3 b.

The discriminator module can be used in two ways:

-   -   either it can be used directly to replace a known discriminator;    -   or else it can be integrated in a known discriminator in order        to provide open loop correction signals to reduce the induced        multipath error.

FIG. 4 is a block diagram of a DLL branch including a discriminatormodule of the invention.

Comparing FIGS. 2 and 4, it can be seen that:

-   -   the branch which generated z_(k) in FIG. 2 is replaced by a        branch which generates z_(k) ⁻⁻ with a delay element of duration        2T_(c);    -   the error signal e_(k) is calculated as a function of z_(k) in        the circuit DISCR using formula (3-12) corresponding to the        module of the invention.

For a signal of SRC type, the S-shaped curve n(ε) is given by:

$\begin{matrix}{{{n(ɛ)}{{\kappa\beta} \cdot {\mathcal{R}( {\frac{g( {( {ɛ - t_{1}} )T_{c}} )}{g( {( {ɛ - t_{2}} )T_{c}} )} + S_{\beta}} )}}}{with}{{g( {ɛ\; T_{c}} )} = {\frac{\sin( {\pi\; ɛ} )}{\pi\; ɛ} \cdot \frac{\cos( {\pi\;\beta\; ɛ} )}{1 - ( {2\;\beta\; ɛ} )^{2}}}}} & ( {3\text{-}15} )\end{matrix}$with for example t1=1.5 and t2=0.5

The discriminator module corresponds to the desired behavior to within agood approximation, i.e. the output of the S-shaped curve isproportional to the input e (e=K.ε) for −0.5ε≦0.5.

By selecting K_(β) in application of formula (3-14) K=1 and e=ε.

The discriminator module can be used to perform open loop estimation tocorrect the code phase output of a conventional module, as shown in FIG.4.

Compared with the FIG. 4 module, there is an additional branch having adelay element 2Tc in order to generate the signal Z_(K) ⁻⁻.

A digital filter H_(com) ^(d)(z) can be used as the lowpass filter atthe output from the new discriminator module. The operation of the DLLbranch remains unchanged compared to the case shown in FIG. 2.

Code phase is corrected by the output from the new discriminator modulewhich is fed through the lowpass filter H_(corr) ^(d)(z). Since the newmodule is less affected by multiple paths, the error contained in thecode phase estimated in the DLL branch can be corrected to a largeextent.

In a variant, the formula can be replaced by an amplitude function:

$e_{k} = {K_{\beta}( {\lambda - \frac{Z_{k}^{--}}{Z_{k}^{--}} + S_{\beta}} )}$where λ is a non-zero number lying between −1 and +1.

In another variant, the error signal e_(k) may be determined by:

$e_{k} = {K_{\beta} \cdot {\mathcal{R}( {\frac{2 \cdot z_{k}^{--}}{z_{k}^{-} + z_{k}^{+}} + S_{\beta}} )}}$

This expression is particularly suitable for open loop correction (FIG.5). Because of the effect of averaging the outputs from the lowpassfilters (Z_(k) ⁻+Z_(k) ⁺)/2, the noise power in the resulting variableis effectively halved, thus leading to less noise in the signal e_(k).

It should be observed that the sampling instants t₁=1.5 and t₂=0.5 canhave other values, and the difference t₁−t₂ between these samplinginstants could be other than one bit or “chip”.

1. A multipath discriminator module for a communications and/ornavigation system that implements spread spectrum modulation, whereinthe module has an input suitable for receiving navigation signals, asampler for supplying sampled signals at a frequency twice the apparentfrequency (fc) of a code of said navigation signals, and a submodule forcalculating an error signal (e_(k)) from said sampled signals(r_(k+1/2)), and a locally generated spreading code (C_(|K|L)) wherein:${e_{k} = {K_{\beta}{\mathcal{R}( {\frac{Z_{\kappa}--}{Z_{\kappa} -} + {S\;\beta}} )}}}\mspace{31mu}$${with}\mspace{20mu}\{ {\begin{matrix}{Z_{\kappa}^{-} = {\lbrack {r_{{k \cdot t}\; 2} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{\kappa}^{--} = {\lbrack {r_{{k \cdot t}\; 1} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}\mspace{25mu}{or}\mspace{20mu}\{ \begin{matrix}{Z_{\kappa}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{\kappa}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} } $ “K_(β)”=constant “h_(k) ^(b)” designatingthe impulse response of a lowpass filter “Z_(K)” designating a signalsample “

(x)” designates a real portion of an operand x${{{and}\mspace{14mu} S_{\beta}} = {- \frac{g( {{- t}\; 1{Tc}} )}{g( {{- t}\; 2{Tc}} )}}}{\mspace{34mu}\mspace{11mu}}$$\;{{{with}\mspace{25mu}{g({aTc})}} = \frac{{Sin}\;\Pi\; a\;{Cos}\;{\Pi\beta}\; a}{\pi\;{a\lbrack {1 - ( {2{\beta a}} )} \rbrack}}}$“β” designating the attenuation factor of the SRC signal “Sβ” is anoffset “Tc” designating a duration of one bit of a pseudo-noise sequence“t1” is a first sampling instant and “t2” is a second sampling instant.2. A multipath discriminator module for a communications and/ornavigation system that implements spread spectrum modulation, whereinthe module has an input suitable for receiving navigation signals, asampler for supplying sampled signals at a frequency twice the apparentfrequency (fc) of a code of said navigation signals, and a submodule forcalculating an error signal (e_(k)) from said sampled signals(r_(k+1/2)), and a locally generated spreading code (C_(|K|L)) wherein:$e_{k} = {K_{\beta}( {\lambda - \frac{Z_{\kappa}--}{Z_{\kappa} -} + {S\;\beta}} )}$“λ” lying in the range −1 to +1 $\begin{matrix}{{with}\mspace{14mu}\{ \begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k - {t\; 2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k - {t\; 1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} } \\{{or}\mspace{14mu}\{ \begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{k}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t\; 1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} }\end{matrix}$ “K_(β)”=constant “h_(k) ^(b)” designating the impulseresponse of a lowpass filter “Z_(K)” designating a signal sample$\begin{matrix}{{{and}\mspace{14mu} S_{\beta}} = {- \frac{g( {{- t}\; 1\mspace{11mu}{Tc}} )}{g( {{- t}\; 2\;{Tc}} )}}} \\{{{with}\mspace{14mu}{g({aTc})}} = \frac{{Sin}\mspace{11mu}\Pi\; a\mspace{11mu}{Cos}\mspace{11mu}\Pi\;\beta\; a}{\pi\;{a\lbrack {1 - ( {2\;\beta\; a} )} \rbrack}}}\end{matrix}$ “β” designating the attenuation factor of the SRC signal“Sβ” is an offset “Tc” designating a duration of one bit of apseudo-noise sequence “t1” is a first sampling instant and “t2” is asecond sampling instant.
 3. A multipath discriminator module for acommunications and/or navigation system that implements spread spectrummodulation, wherein the module has an input suitable for receivingnavigation signals, a sampler for supplying sampled signals at afrequency twice the apparent frequency fc of a code of said navigationsignals, and a submodule for calculating an error signal e_(k) from saidsampled signals (r_(k+1/2)), and a locally generated spreading code(C_(|K|L)) wherein: $\begin{matrix}{e_{k} = {K_{\beta}\mspace{14mu}( {{2\frac{Z_{K}^{--}}{Z_{K}^{-} + Z_{K}^{+}}} + S_{\beta}} )}} \\{{with}\mspace{14mu}\{ \begin{matrix}{Z_{K}^{+} = {\lfloor {r_{k + 0.5} \cdot C_{{K}L}} \rfloor \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k - {t\; 2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{{k\; t} - {t\; 1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} } \\{{or}\mspace{14mu}\{ \begin{matrix}{Z_{K}^{+} = {\lbrack {r_{k} \cdot C_{{{k + {1/2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{({k - {t\; 2}})}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{k}^{--} = {\lbrack {r_{k} \cdot C_{{({k - {t\; 1}})}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} }\end{matrix}$ “

(x)” designates a real portion of an operand x “K_(β)”=constant “h_(k)^(b)” designating the impulse response of a lowpass filter “Z_(K)”designating a signal sample $\begin{matrix}{{{and}\mspace{14mu} S_{\beta}} = {- \frac{g( {{- t}\; 1\mspace{11mu}{Tc}} )}{g( {{- t}\; 2\;{Tc}} )}}} \\{{{with}\mspace{14mu}{g({aTc})}} = \frac{{Sin}\mspace{11mu}\Pi\; a\mspace{11mu}{Cos}\mspace{11mu}\Pi\;\beta\; a}{\pi\;{a\lbrack {1 - ( {2\;\beta\; a} )} \rbrack}}}\end{matrix}$ “β” designating the attenuation factor of the SRC signal“Sβ” is an offset “Tc” designating a duration of one bit of apseudo-noise sequence “t1” is a first sampling instant and “t2” is asecond sampling instant.
 4. A discriminator module according to claim 1,wherein:${K\;\beta} = \frac{1}{\frac{\mathbb{d}\;}{\mathbb{d}ɛ}( \frac{g( {( {ɛ - {t\; 1}} ){Tc}} )}{g( {( {ɛ - {t\; 2}} ){Tc}} )} )_{ɛ = 0}}$ε designating the code phase error.
 5. A discriminator module accordingto claim 1, wherein t1−t2=1.
 6. A discriminator module according toclaim 5, wherein t1=1.5 and t2=0.5.
 7. A navigation system, presenting adiscriminator module according to claim
 1. 8. A navigation system,presenting a discriminator module generating an error signal e′_(k)serving in conventional manner to correct a closed loop on the basis ofsampled signal Z+_(K) and Z−_(K) which is associated with adiscriminator module according to claim 1 in order to generate in anopen loop a correction signal for a code phase output.